Circuits and methods for eliminating reference spurs in fractional-n frequency synthesis

ABSTRACT

Disclosed are circuits and method for reducing or eliminating reference spurs in frequency synthesizers. In some implementations, a phase-locked loop (PLL) such as a Frac-N PLL of a frequency synthesizer can include a phase frequency detector (PFD) configured to receive a reference signal and a feedback signal. The PFD can be configured to generate a first signal representative of a phase difference between the reference signal and the feedback signal. The PLL can further include a compensation circuit configured to generate a compensation signal based on the first signal. The PLL can further includes a voltage-controlled oscillator (VCO) configured to generate an output signal based on the compensation signal. The compensation signal can include at least one feature for substantially eliminating one or more reference spurs associated with the PLL.

CROSS-REFERENCE TO RELATED APPLICATIONS

Any and all applications for which a foreign or domestic priority claimis identified in the Application Data Sheet as filed with the presentapplication are hereby incorporated by reference under 37 CFR 1.57.

BACKGROUND

Field

The present disclosure generally relates to circuits and method forreducing or eliminating reference spurs in fractional-N frequencysynthesis in radio-frequency (RF) applications.

Description of the Related Art

A radio-frequency (RF) communication system typically relies on one ormore operating frequencies. A clean and stable signal having a desiredoperating frequency is highly desirable.

In many RF systems, a frequency synthesizer can provide a signal with anoperating frequency through use of a phase-locked loop (PLL). In manyoperating situations, a PLL can suffer from undesirable effects such asphase noise or spurious tones (spurs). Phase noise typically refers tosmall variations from the intended operating frequency. Spurs aretypically large unwanted frequency components that manifest themselvesat some offset of the operating frequency.

SUMMARY

In some implementations, the present disclosure relates to aphase-locked loop (PLL) circuit for a frequency synthesizer of awireless device. The PLL circuit includes a phase frequency detector(PFD) configured to receive a reference signal and a feedback signal.The PFD is further configured to generate a first signal representativeof a phase difference between the reference signal and the feedbacksignal. The PLL further includes a compensation circuit in communicationwith the PFD. The compensation circuit is configured to generate acompensation signal based on the first signal. The PLL further includesa voltage-controlled oscillator (VCO) in communication with thecompensation circuit. The VCO is configured to generate an output signalbased on the compensation signal. The compensation signal includes atleast one feature for substantially eliminating one or more referencespurs associated with the PLL.

In some embodiments, the compensation circuit can include a charge pumpconfigured to receive the first signal and generate a current signal tocompensate for the phase difference. The compensation circuit canfurther include a loop filter configured to receive the current signaland generate a corresponding voltage signal, with the voltage signalbeing provided to the VCO.

In some embodiments, the at least one feature of the compensation signalcan include the current signal having a pulse with a substantiallyconstant width and an amplitude representative of the phase difference.The PFD can be configured to generate an up signal or a dn as the firstsignal, with the up signal being generated when a phase of the referencesignal leads a phase of the feedback signal, and the dn signal beinggenerated when a phase of the reference signal lags a phase of thefeedback signal. The compensation circuit can include a charging circuitin communication with the PFD. The charging circuit can be configured tocharge a capacitance element upon receipt of the first signal, with thecharged capacitance having a voltage representative of the phasedifference. The charging circuit can include a charging switchconfigured to facilitate the charging of the capacitance element, and adrain switch configured to facilitate draining of the charge in thecapacitance element.

In some embodiments, the compensation circuit can further include avoltage-to-current converter in communication with the charging circuit.The voltage-to-current converter can be configured to generate a controlsignal in response to the voltage of the charged capacitance element.The charge pump can be in communication with the voltage-to-currentconverter, with the charge pump being configured to generate the currentsignal based on the control signal from the voltage-to-currentconverter. The charge pump can include a first current source configuredto generate a positive current as the current signal for the up signaland a second current source configured to generate a negative current asthe current signal for the dn signal. The current signal can bemodulated based on a combination of the up signal and a pulse signal ora combination of the dn signal and the pulse signal.

In some embodiments, the compensation circuit can further include afirst switch in communication with the first current source and a secondswitch in communication with the second current source, with the firstand second switches being configured to be controlled to provide themodulation of the current signal. In some embodiments, the first switchor the second switch can be engaged only when the capacitance elementhas a substantially full charge. In some embodiments, the chargingswitch is not engaged if either of the first switch and the secondswitch is engaged.

In some embodiments, the compensation circuit can further include afirst control block in communication with the first switch and a secondcontrol block in communication with the second switch. The first controlblock can be configured to generate a first enable pulse for the firstswitch based on a combination of the up signal and the pulse signal. Thesecond control block can be configured to generate a second enable pulsefor the second switch based on a combination of the dn signal and thepulse signal. Each of the first and second control blocks can beconfigured to generate in internal signal that goes high with a risingedge of a respective one of the up and dn signals. The internal signalcan be further configured to go low with a falling edge of the pulsesignal. Each of the first and second control blocks can be furtherconfigured to combine the internal signal with the pulse signal to yielda respective one of the first and second enable signals. Each of thefirst and second control blocks can be configured to perform an ANDoperation to combine the internal signal with the pulse signal. Thepulse signal can include a plurality of pulses, with each pulse having asubstantially constant width and going high with a falling edge of thereference signal.

In some embodiments, the PLL circuit can further includes a dividercircuit in communication with the VCO and the PFD. The divider circuitcan be configured to receive the output signal from the VCO and generatean updated version of the feedback signal. The PLL can be a Frac-N PLL.The PLL can further include a sigma delta modulator (SDM) incommunication with the divider circuit to form a loop. The loop can beconfigured to allow the output signal to have an output frequency thatis a non-integer multiple of the frequency of the reference signal.

In accordance with a number of implementations, the present disclosurerelates to a method for operating a phase-locked loop (PLL) of afrequency synthesizer in a wireless device. The method includesreceiving a reference signal and a feedback signal. The method furtherincluded generating a first signal representative of a phase differencebetween the reference signal and the feedback signal. The method furtherincludes generating a compensation signal based on the first signal. Themethod further includes generating an output signal based on thecompensation signal. The compensation signal includes at least onefeature for substantially eliminating one or more reference spursassociated with the PLL.

In a number of implementations, the present disclosure relates to aradio-frequency (RF) module that includes a packaging substrateconfigured to receive a plurality of components. The module furtherincludes one or more die mounted on the packaging substrate. The one ormore die include a frequency synthesizer circuit, with the frequencysynthesizer circuit having a phase-locked loop (PLL) circuit. The PLLincludes a phase frequency detector (PFD) configured to receive areference signal and a feedback signal. The PFD is further configured togenerate a first signal representative of a phase difference between thereference signal and the feedback signal. The PLL further includes acompensation circuit in communication with the PFD. The compensationcircuit is configured to generate a compensation signal based on thefirst signal. The PLL further includes a voltage-controlled oscillator(VCO) in communication with the compensation circuit. The VCO isconfigured to generate an output signal based on the compensationsignal. The compensation signal includes at least one feature forsubstantially eliminating one or more reference spurs associated withthe PLL.

In some implementations, the present disclosure relates to a wirelessdevice that includes an antenna configured to facilitate reception of aradio-frequency (RF) signal. The wireless device further includes areceiver in communication with the antenna, with the receiver beingconfigured to process the RF signal. The wireless device furtherincludes a frequency synthesizer in communication with the receiver,with the frequency synthesizer circuit having a phase-locked loop (PLL)circuit. The PLL includes a phase frequency detector (PFD) configured toreceive a reference signal and a feedback signal. The PFD is furtherconfigured to generate a first signal representative of a phasedifference between the reference signal and the feedback signal. The PLLfurther includes a compensation circuit in communication with the PFD.The compensation circuit is configured to generate a compensation signalbased on the first signal. The PLL further includes a voltage-controlledoscillator (VCO) in communication with the compensation circuit. The VCOis configured to generate an output signal based on the compensationsignal. The compensation signal includes at least one feature forsubstantially eliminating one or more reference spurs associated withthe PLL.

In some embodiments, the wireless device can further include atransmitter in communication with the antenna. The transmitter can beconfigured to generate a transmit signal.

For purposes of summarizing the disclosure, certain aspects, advantagesand novel features of the inventions have been described herein. It isto be understood that not necessarily all such advantages may beachieved in accordance with any particular embodiment of the invention.Thus, the invention may be embodied or carried out in a manner thatachieves or optimizes one advantage or group of advantages as taughtherein without necessarily achieving other advantages as may be taughtor suggested herein.

BRIEF DESCRIPTION OF THE DRAWINGS

FIG. 1 schematically depicts a phase-locked loop (PLL) having a spurremoval component.

FIG. 2 shows a wireless device in which the PLL of FIG. 1 can beimplemented.

FIG. 3 shows that in some embodiments, the PLL of FIG. 1 can beimplemented in a frequency synthesizer that facilitates processing of areceived radio-frequency (RF) signal.

FIG. 4 shows an example PLL that is capable of operating as a Frac-NPLL.

FIGS. 5A and 5B show an example charge pump that can be implemented inthe PLL of FIG. 4, and an example operating configuration of the chargepump.

FIGS. 6A and 6B show examples of a loop filter that can be implementedin the PLL of FIG. 4.

FIG. 7 shows an example characteristics of a voltage controlledoscillator (VCO) that can be implemented in the PLL of FIG. 4.

FIG. 8 shows an example divider that can be implemented in the PLL ofFIG. 4.

FIG. 9 shows an example operating configuration of a sigma deltamodulator (SDM) that can be implemented in the PLL of FIG. 4 to allowthe PLL to operate as a Frac-N.

FIG. 10 shows an example charge pump current that can be implemented inthe PLL FIG. 4.

FIG. 11 shows an example result of simulation to demonstrate one or moreeffects of a charge pump pulse on reference spurs.

FIG. 12 shows another example result of simulation to demonstrate one ormore effects of a charge pump pulse on reference spurs.

FIG. 13 shows yet another example result of simulation to demonstrateone or more effects of a charge pump pulse on reference spurs.

FIG. 14 shows an example of a fixed-width variable-amplitude (FWVA)pulse that can be implemented to facilitate removal of reference spurs.

FIG. 15 shows another example of an FWVA pulse signal.

FIG. 16 shows a frequency domain representation of the time domainsignal of FIG. 15.

FIG. 17 shows an example configuration of a fixed width variableamplitude charge pump (FWVACP).

FIG. 18 shows timing diagrams for various parameters associated with theFWVACP configuration of FIG. 17.

DETAILED DESCRIPTION OF SOME EMBODIMENTS

The headings provided herein, if any, are for convenience only and donot necessarily affect the scope or meaning of the claimed invention.

In some embodiments, a radio-frequency (RF) device such as a wirelessdevice can include a frequency synthesizer having a phase-locked loop(PLL). FIG. 1 schematically depicts a PLL 100 that can be configured toreceive a reference signal and generate an output signal having adesired output frequency. Such a PLL can include a spur removalcomponent having one or more desirable features as described herein.Such a spur removal component is sometimes described herein as acompensation circuit, a compensation component, and the like.

In some embodiments, a PLL having one or more features of the presentdisclosure can be implemented in a radio-frequency (RF) device such as awireless device. Such a wireless device can include, for example, acellular phone, a smart-phone, a hand-held wireless device with orwithout phone functionality, a wireless tablet, etc. Although describedin the context of a wireless device, it will be understood that one ormore features of the present disclosure can also be implemented in otherRF systems, including, for example, a base-station.

FIG. 2 schematically depicts an example wireless device 110 having oneor more advantageous features described herein. The wireless device 110is shown to include an antenna 140 configured to facilitate transmission(Tx) and/or reception (Rx) of RF signals. Such Tx and/or Rx operationscan be performed simultaneously by use of a duplexer 138. Althoughdescribed in the context of such duplex functionality and commonantenna, other configurations are also possible.

A received signal is shown to be routed from the antenna 140 to areceiver circuit 120 via the duplexer 138 and a low-noise amplifier(LNA) 130. For transmission, a signal to be transmitted is shown to begenerated by a transmitter circuit 126 and routed to the antenna 140 viaa power amplifier (PA) 136 and the duplexer 118. The receiver circuit120 and the transmitter circuit 126 may or may not be part of a samecomponent (e.g., a transceiver). In some embodiments, a wireless device110 can include both of the receiver and transmitter circuits, or justone circuit (e.g., receiver or transmitter).

The wireless device 110 is shown to further include a frequencysynthesizer circuit 122 having a phase-locked loop (PLL) 100. Such acircuit (122) can include one or more features as described herein toprovide advantages for either or both of Rx and Tx functionalitiesassociated with the wireless device 110.

The receiver circuit 120, the transmitter circuit 126, and the frequencysynthesizer circuit 122 are shown to be in communication with a basebandsubsystem 114 which can include, for example, a processor 116 configuredto control a number of operations associated with the wireless device110, and a memory 118 configured to store data, executable instructions,etc. The baseband subsystem 114 is also shown to be in communicationwith a user interface 112 to allow interfacing of variousfunctionalities of the wireless device 110 with a user.

As shown in FIG. 2, at least some of the one or more features associatedwith the frequency synthesizer 122 can be implemented in an RF module102. Such a module can include a packaging substrate configured toreceive a plurality of components. The module 102 can include one ormore semiconductor die mounted on the packaging substrate. Such one ormore die can include some or all of the circuit that provides variousfunctionalities associated with the frequency synthesizer 122.

FIG. 3 shows an example configuration 150 where one or more frequencysynthesizers can be implemented in a receiver chain of a wirelessdevice. Although described in such a receiver chain context, it will beunderstood that one or more features of the present disclosure can alsobe implemented in other parts of a wireless device.

A signal received by the antenna 140 can be passed through a preselectfilter 152 configured to pass a desired receive band. The preselectfilter 152 can work in conjunction with an image filter 156 to furtherisolate the receive band. Both of these filters can pass substantiallythe entire receive band, since channel selection does not occur untilmore downstream of the receiver chain.

A low-noise amplifier (LNA) 130 can be implemented to boost the incomingsignal. Such an LNA can be configured to provide this gain whiledegrading the signal-to-noise ratio (SNR) as little as possible. Anautomatic gain control (AGC) circuit 154 can be configured to allow thewireless device to handle a wide range of expected input power levels.For example, a low powered incoming signal can require a greater boostthan a higher powered incoming signal.

A first mixer 158 a can be configured to convert the RF channels down tolower frequencies and center a desired channel at a specificintermediate frequency (IF). Such a specific IF can be provided to thefirst mixer 158 a from a first frequency synthesizer 122 a.

At this stage, the entire received-and-filtered band is now mixed downto the IF. An IF filter 160 can be configured to isolate the channel ofinterest from the receive band. An AGC circuit 162 can be configured toallow the wireless device to handle a wide range of expected input powerlevels associated with the isolated channel of interest.

A second mixer 158 b can be configured to convert the foregoing isolatedchannel signal down to a baseband signal. Such down-conversion can befacilitated by a second frequency synthesizer 122 b configured togenerate and provide a desired baseband frequency to the second mixer158 b.

An AGC circuit 164 can be configured to allow the wireless device tohandle a wide range of expected input power levels associated with theoutput of the second mixer 158 b. A baseband filter 166 can beconfigured to filter the selected baseband-frequency signal beforehaving the signal sampled by an analog-to-digital converter (ADC) 168. Adigital signal resulting from such an ADC can be passed to a basebandsub-system (not shown in FIG. 3).

In the context of the example signal processing configuration of FIG. 3,the first frequency synthesizer 122 a generates a clock signal thatfacilitates the down-conversion of a received signal to an IF signal.Similarly, the second frequency synthesizer 122 b generates a clocksignal that facilitates the down-conversion of the IF signal to abaseband signal.

As described in reference to FIGS. 1 and 2, a frequency synthesizer caninclude a PLL. In some embodiments, a PLL can be implemented as anegative feedback control system designed to generate an output at aparticular frequency. Such an output can be utilized as an output of thefrequency synthesizer.

FIG. 4 shows an example configuration of a PLL circuit 100 which can bea part of a frequency synthesizer 122. As shown, an input 170 into thePLL 100 is a reference signal which is typically provided from a crystaloscillator (not shown) to a phase frequency detector (PFD) 172. In someembodiments, the PFD 172 can be configured to compare the rising edgesof the reference signal and a feedback signal (in path 196) anddetermine if the feedback signal is leading or lagging with respect tothe reference signal. Based on this comparison, the PFD 172 can output asignal to a charge pump 176 (through path 174). In response, the chargepump 176 can output a current which is related to the phase differencebetween the reference and feedback signals.

The foregoing charge pump current can be provided to a loop filter 180(through path 178). The loop filter 180 can be configured to convert thecharge pump current into a voltage suitable for driving a voltagecontrolled oscillator (VCO) 184 (through path 182). The loop filter 180can also be configured to control loop dynamics of the PLL (e.g.,bandwidth, settling time, etc.).

The VCO 184 can be configured to output a signal (through path 186)having a frequency that is related to the driving voltage from the loopfilter 180. In some embodiments, such an output of the VCO 184 can alsobe an output of the PLL 100 (through path 188). Accordingly, VCO outputand PLL output are sometimes used interchangeably in the descriptionherein.

The output of the VCO 184 can be fed into a divider circuit 192 (throughpath 192). The divider circuit 192 can be configured to divide the VCOoutput frequency back down to the reference frequency. A feedback signalfrom the divider circuit 192 can be fed back into the PFD 172 (throughpaths 194, 196) to thereby complete the PLL loop.

The foregoing feedback mechanism allows the output frequency of the PLLto lock on to a frequency that is a multiple of the reference signalfrequency. If the multiple is an integer, the PLL is considered to be anInteger-N PLL. If the multiple contains a fractional component, the PLLis considered to be a Frac-N PLL (or Fractional-N PLL).

FIG. 4 further shows a sigma delta modulator (SDM) 200 in communicationwith the above-described feedback loop. As described herein, such an SDMcan be configured as an additional feedback loop with the dividercircuit 192 (through paths 194, 198 from the divider circuit 192 to theSDM 200, and through path 202 from the SDM 200 to the divider circuit192) to allow the PLL 100 to operate as a Frac-N PLL.

In some embodiments, the SDM 200 can be configured to generate a signalthat instructs the divider circuit 192 with which integer value todivide the frequency of the VCO output signal. By way of an example,suppose that a PLL has a reference signal frequency of 40 MHz, and it isdesired to output a signal having a frequency of 2.41 GHz. Such aconfiguration yields a divide ratio of 60 and ¼. One way the PLL canachieve this divide ratio is to implement dividing by 60 for threereference cycles, then dividing by 61 for one cycle. This pattern canthen repeat. Over each repetition the average divide value, N_(avg), is60 and ¼ as expected.

In the context of the foregoing example, the SDM 200 can instruct thedivider circuit 192 to divide by 60 or 61. Such dithering between twointeger divide ratios can allow the divider circuit 192 to beimplemented even if the circuit (192) is only capable of integerdivision. Accordingly, the output frequency of such a Frac-N PLL can bean averaged result of a plurality of integer divide values.

It is noted that the output resolution can be dependent on the divideratio and not on the reference clock. In another example, now supposethat the desired output frequency is 2.405 GHz which has a greaterresolution than the foregoing example of 2.41 GHz. Such a frequency canbe synthesized by changing the divide ratio from 60 and ¼ to 60 and ⅛.

FIGS. 5-9 show example configurations and/or characteristics of thevarious components of the PLL 100 of FIG. 4. FIG. 5A schematicallydepicts an example configuration where the PFD 172 is working in concertwith the charge pump 176 to produce pulses of current (i_(CP)) which areproportional to the phase difference between the reference signal (170)and the feedback signal (196).

The pulses of current, i_(CP), generated by the charge pump 176 (to path178), can alter the tuning voltage of the VCO (not shown in FIG. 5A).Consequently, the phase of the feedback signal emerging from the VCO canbe either increased or decreased to more closely match the referencephase.

FIG. 5B shows an example of how the PFD 172 and the charge pump 176 canoperate. At time A the feedback signal is shown to go high before thereference signal. This can indicate that the feedback signal is leadingthe reference signal. For such a situation, the VCO output frequency canbe decreased, to effectively delay the feedback signal. To accomplishsuch a response, the PFD 172 can issue a DOWN pulse (224) until time Bat which the reference signal has its rising edge. Until the next risingedge (e.g., at time C) of either the reference or feedback signals, thePFD 172 can become inactive. At time C, since both rising edges of thereference and feedback signals occur substantially simultaneously, thereis no UP or DOWN signal being issued by the PFD 172. At time D, thereference signal is now depicted as leading the feedback signal.Accordingly, an UP pulse (222) can be issued by the PFD 172 during aperiod between times D and E to thereby increase the VCO outputfrequency.

FIGS. 6A and 6B show examples of filters that can be implemented for theloop filter 180 described in reference to FIG. 4. In some embodiments,such a loop filter can be configured to provide a plurality offunctionalities. For example, the loop filter 180 can be configured toconvert the charge pump current (e.g., i_(CP) in FIG. 5A) into a voltagesuitable for driving the VCO (184 in FIG. 4). In another example, theloop filter can be configured to control the loop dynamics of the PLL100 (e.g., bandwidth, settling time, etc.).

FIG. 6A shows that in some embodiments, a second-order passive filtercan be implemented in the loop filter 180. Such a passive filter caninclude first and second paths to a ground from a signal path 230. Thefirst path can include resistance R and capacitance C1 connected inseries between the signal path 230 and the ground. The second path caninclude capacitance C2 between the signal path 230 and the ground.

FIG. 6B shows that in some embodiments, a third-order passive filter canbe implemented in the loop filter 180. Such a passive filter can includefirst, second, and third paths to a ground from a signal path 230. Thefirst path can include resistance R and capacitance C1 connected inseries between the signal path 230 and the ground. The second path caninclude capacitance C2 between the signal path 230 and the ground. Thethird path can include capacitance C3 between the signal path 230 andthe ground.

FIG. 7 shows an example response that the VCO (184) of FIG. 4 can beconfigured to generate. In some embodiments, a VCO can be an oscillatorwhose output frequency is related to its input voltage. FIG. 7 showsthat in some embodiments, such voltage-dependence of the outputfrequency can be a linear or approximately linear relationship. One cansee that as the tuning voltage (v_(Tune)) increases, the outputfrequency increases proportionally within a given range. For example, aVCO can have a range of voltages over which it can operate. This rangeis shown as V_(Min) to V_(Max), with V_(Nom) being a nominal voltage atwhich the VCO operates.

As further shown in FIG. 7, a slope parameter k_(vco) can be expressedin units of Hz/V. In some situations, it can also be expressed inradians/Sec/V. In some embodiments, a VCO can be configured so that withno input voltage to the VCO (e.g., v_(Tune)=0), the output is thenominal frequency, f_(nom). The control voltage v_(Tune) can be appliedto move the output frequency away from the nominal frequency. Forexample, an instantaneous frequency (f_(vco)) of the VCO can begenerated as f_(vco)=f_(nom) k_(vco)v_(tune).

FIG. 8 depicts an example of how the divider circuit 192 of FIG. 4 canbe configured to reduce the frequency of an input signal 240 (throughinput path 190) to yield an output signal 242 (through output path 194)having a reduced frequency. In some embodiments, as described herein,the divider circuit 192 can be configured to receive an output of theVCO (184 in FIG. 4) and divide the frequency of the VCO-output signal bywhatever integer it is programmed with. As also described herein,non-integer division values of the divider circuit 192 can also beimplemented with use of the SDM (200 in FIG. 4) so as to facilitate aFrac-N PLL configuration.

As described herein, an SDM (200 in FIG. 4) can be incorporated into aPLL to allow the PLL to operate as a Frac-N PLL. FIG. 9 shows an exampleof how an SDM can be configured to provide different SDM orders.

As described herein, an SDM can allow the PLL to output a signal thatrepresents a fractional portion of the divide value. FIG. 9 shows thatdifferent orders can be implemented for SDMs to yield such afractional-divided output. The examples shown in FIG. 9 are in thecontext of an example divide value fraction of ¼; however, it will beunderstood that divisions by other fractions can also be implemented. Itwill also be understood that, for a given order, other SDM architecturescan be implemented.

In FIG. 9, a clock trace is shown at the top. A 1st order SDM output isdepicted as having a high for one clock cycle and low for the nextthree. Such a pattern can repeat. The length of this pattern in terms ofclock cycle is commonly referred to as an SDM sequence length. In theforegoing 1st order SDM example, the SDM sequence length is 4.

In some embodiments, 2nd and 3rd order SDM configurations can have amore complicated patterns. However, in the context of the ¼ fractionexample, such 2nd and 3rd order configurations still average out to ¼over their respective sequence lengths. It is also noted that thesequence length may or may not increase as the order of the SDMincreases.

As described herein by way of examples, various features associated withthe feedback mechanism can allow the output frequency of the PLL to lockon to a desired frequency that is a non-integer multiple of thereference signal. As also described herein, such a non-integer multiplefunctionality can be facilitate by an integer dithering technique.

In some situations, such an integer dithering operation in a PLL canyield a side-effect that includes spurious tones, or spurs. A spur istypically considered to be an unwanted tone in the frequency domain at aparticular frequency.

Spurs can be large and can appear as distinct spikes that rise above anoise floor. In some situations, spurs can be reference spurs thattypically appear in the frequency domain at or near multiples of thereference frequency. Such spurs can occur in either Integer-N or Frac-NPLLs. Frac-N spurs typically can result from an SDM switching theinteger divider value of the PLL.

In some situations, reference spurs can arise from the followingoperating conditions. In an ideal Integer-N PLL that has settled, thecharge pump will output zero current as the loop filter holds theselected voltage for the VCO to thereby output the desired frequency.Charge pump mismatch can occur when rising edges of the feedback andreference signal occur simultaneously. For example, the PFD can outputboth UP and DOWN pulses until the PFD can reset itself. If the UP andDOWN pulse are mismatched, there can be an unwanted charge put on theloop filter that needs to be compensated for. Such compensation canoccur periodically with the reference frequency, and can be asignificant cause of reference spurs in an Integer-N PLL.

In some implementations, a Frac-N PLL is not truly settled because ofthe dithering provided by the SDM. Accordingly, current from the chargepump also will not settle to zero, although it can average toapproximately zero. Instead, the charge pump pulse can exhibit violentchanges in response to the SDM. Charge pump mismatch can also be anissue. However, as described herein, reference spurs can still beproduced from an ideal charge pump with no mismatch.

In some implementations, a charge pump pulse can be utilized to generatenulls in the PLL output signal. To facilitate such a technique, one canconsider how frequency components in the charge pump current can appearat the output of the PLL. Frequency components can first be attenuatedby the loop filter. No mixing occurs at this point, so frequency nullsin the charge pump can still exist in the voltage signal at the VCOinput. Even with the assumption that the VCO is linear, there still canexist a possibility for mixing of frequency components due to widebandFM modulation. If the modulation index of each frequency component atthe VCO input is small enough, only narrowband FM modulation will occur.Since narrowband FM is generally linear, no new frequency componentswill be generated. Therefore frequency components that were nulled inthe charge pump signal will still be nulled at the PLL output. Usingsuch reasoning, one can implement a technique for reducing oreliminating particular frequencies in the charge pump current, andtherefore the PLL output.

FIG. 10 shows an example charge pump current (i_(ce)) found in a Frac-NPLL, such as the example described in reference to FIG. 4. Examples ofreference and feedback signals, as well as example reference edgemarkers 250 a, 250 b are shown relative to the charge pump currenttrace.

In some implementations, the charge pump current pulse can be affectedin two ways by the reference and feedback signals. First, the width ofthe charge pump pulse can dependent upon the spacing between the risingedges of the two signals. This is sometimes called pulse widthmodulation. Second, the charge pump pulse may occur either before orafter the reference rising edge, depending on whether the referencesignal leads or lags the feedback signal. This is sometimes called pulseposition modulation. In some situations, these two factors can cause thecharge pump to generate reference spurs. As described herein, otherconditions such as charge pump mismatch can exacerbate reference spurs.

FIGS. 11-13 show example results of various simulations to demonstrateone or more effects of the charge pump pulse in terms of referencespurs. In FIG. 11, a simple pulse train that has neither pulse width norpulse position modulation is shown. Pulse trains corresponding to panels(a) to (c) are time domain signals; and their corresponding frequencydomain signals are shown in panels (d) to (f).

The time domain rectangular pulses of panels (a) and (b) are representedby sync functions in the frequency domain (panels (d) and (e)). For thepurpose of description herein, “sync” function can be usedinterchangeably with “sinc” function to refer to (sin(x)/x). In theexample of FIG. 11, both of these two pulses have a frequency of 10 MHz.Such a frequency determines the spacing of the spurs as shown in panels(d) and (e). The example pulses have a 10% duty cycle. This leads to thefirst sync function null 262 occurring at 100 MHz. In someimplementations, it is possible to change the location of this null bychanging the pulse width. For example, larger pulse widths can move thenull to lower frequencies. One or more features associated with thistechnique can be utilized to reduce or eradicate any frequencycomponents after the first reference frequency by setting the pulsewidth.

The signal train of panel (c) corresponds to a sum of the signal trainsof panels (a) and (b). The rising edges (260) of the pulses now occur at20 MHz, which is analogous to the reference frequency in a PLL. It isnoted is that there is no frequency components at integer multiples of20 MHz. Such a feature is described herein in greater detail.

In FIG. 12, a phase shift is introduced in the signal train of panel(b). Introduction of such a phase shift can simulate pulse positionmodulation. Frequency domain representations of panels (d) and (e) aresimilar to those of panels 11(d) and 11(e), since phase information isnot shown in the frequency domain results. However, when the signaltraces of panels (a) and (b) are summed together, panel (f) shows thatthere are no longer nulls at multiples of the reference frequency. Thatis, reference spurs are generated at frequencies where nulls are in theexample of FIG. 11. This example demonstrates that pulse positionmodulation can contribute to reference spurs.

In FIG. 13, up pulses of the signal train of panel (d) is similar tothat of panel (c) of FIG. 11. Down pulses of the signal train of panel(d), however, have pulse width modulation applied to them. The entiresignal of panel (d) still averages out to zero. As with the pulseposition modulation example of FIG. 12, pulse width modulation generatesreference spurs.

With the examples of FIGS. 11-13, it is shown that both pulse positionmodulation and pulse width modulation can contribute to reference spurs.To reduce or eradicate reference spurs then, a pulse train having thecharacteristics of the pulse in panel (c) of FIG. 11 can be utilized.More specifically, a pulse train from the charge pump can be configuredto have substantially constant width pulses that begin substantially atthe rising edges of the reference signal.

FIG. 13 provides an explanation of how Frac-N spur spacing can berelated to a sequence length of a pulse train. The pulse train of panel(d) has a sequence length of 4. That is, the signal repeats every 4pulses. Based on superposition principle, the signal of panel (d) can bebroken down into the signals of panels (a) to (c). The signal of panel(a) has a period of 100 ns, thereby yielding frequency components with aspacing of 10 MHz. The signals of panels (b) and (c) have a period of200 ns, thereby yielding frequency components with a spacing of 5 MHz.

Based on the foregoing, Frac-N spur location and how it relates to thereference frequency can be expressed as

n _(th) spur location=n f _(ref) /L _(s)  (1)

where L_(s) is the sequence length of the pulse train. From thisequation, it can be seen that there are L_(s)−1 spurs between eachreference frequency. Additionally, every L_(s) ^(th) spur is a referencefrequency.

In some implementations, reference spurs can be reduced or eliminatedfrom a charge pump pulse by elimination of pulse position modulationand/or pulse width modulation. By way of an example, such eliminationcan be achieved by generating a fixed-width variable-amplitude (FWVA)pulse. In some PLLs, pulse position modulation and pulse widthmodulation can be responsible for representing phase error information.With a FWVA pulse, phase error information can be stored in or berepresented by the amplitude of the pulse.

An example FWVA pulse train is shown in FIG. 14. In someimplementations, following restrictions can be placed on the signal. Forexample, the width of each pulse (“Pulse width” in panel (a)) can besubstantially equal. In another example, the spacing between pulses(“Pulse spacing” in panel (a)) can also be substantially equal. In yetanother example, the amplitude of each pulse can be set to be arbitraryto solve a general case, and can be defined as a₁, a₂, . . . , a_(p). Inyet another example, the signal can be made to be substantiallyperiodic.

Discrete Fourier analysis can be utilized to examine the frequencyspectrum of the example signal of FIG. 14. Such an analysis can show asubstantially total cancellation of the frequency components atfrequencies related to the beginning edges of each pulse. Thesefrequencies are analogous to integer multiples of the referencefrequency in a PLL.

In the example of FIG. 14, the signal depicted in panel (a) iscontinuous in the time domain. Such a signal can be sampled for use in adiscrete-time Fourier analysis. Panel (b) of FIG. 14 shows the signalafter it has been sampled.

Based on the sampling of panel (b), one can define a discrete signalx(n) as

x(n){a ₁ a ₁ . . . a ₁ 0 0 . . . 0 0 a ₂ a ₂ . . . a ₂ 0 0 . . . 0 0 a_(p) a _(p) . . . a _(p) 0 0 . . . 0 0}.  (2)

Using a discrete Fourier equation, one can generate the spur amplitudesof the reference frequencies. Consider the following example analysisequation for discrete-time periodic signals:

$\begin{matrix}{{c_{k} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1}\; {{x(n)}^{{- {({{j2\pi}\; {kn}})}}/N}}}}}{c_{k} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1}\; {{x(n)}^{{- {({{j2\pi}\; {kn}})}}/N}}}}}} & (3)\end{matrix}$

where k is the index of the frequency component. Other parameters orvariables associated with FIG. 14 and Equation 3 are listed in Table 1.

TABLE 1 Parameter/Variable Represents x(n) Time domain signal p Totalnumber of pulses per period N n Index of the signal x(n) w Number ofsamples per pulse z Number of zeros between pulses r Reference period:w + z N Period of signal: p(w + z)

By substituting x(n) of Equation 2 into Equation 3, c_(k) can beexpressed as

$\begin{matrix}{{c_{k} = {\frac{1}{N}\left\lbrack {{a_{1}\left( {1 + ^{- \frac{{j2\pi}\; k}{N}} + ^{- \frac{{j2\pi}\; k\; 2}{N}} + \ldots + ^{- \frac{{j2\pi}\; {k{({w - 1})}}}{N}}} \right)} + {a_{2}\left( {^{- \frac{{j2\pi}\; {k{(r)}}}{N}} + ^{- \frac{{j2\pi}\; {k{({r + 1})}}}{N}} + \ldots + ^{- \frac{{j2\pi}\; {k{({r + w - 1})}}}{N}}} \right)} + {a_{3}\left( {^{- \frac{{j2\pi}\; {k{({2\; r})}}}{N}} + ^{- \frac{{j2\pi}\; {k{({{2\; r} + 1})}}}{N}} + \ldots + ^{- \frac{{j2\pi}\; {k{({{2\; r} + w - 1})}}}{N}}} \right)} + \ldots + {a_{p}\left( {^{- \frac{{j2\pi}\; {k{({{({p - 1})}r})}}}{N}} + ^{- \frac{{j2\pi}\; {k{({{{({p - 1})}r} + 1})}}}{N}} + \ldots + ^{- \frac{{j2\pi}\; {k{({{{({p - 1})}r} + w - 1})}}}{N}}} \right)}} \right\rbrack}}\mspace{20mu} {c_{k} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1}\; {{x(n)}^{{- {({{j2\pi}\; {kn}})}}/N}}}}}} & (4)\end{matrix}$

From Equation 1, the number of spurs between each reference frequency isequal to the sequence length, or p in this analysis. Accordingly, thefirst reference spur occurs at c_(p). Solving for c_(p) (using N=pr fromTable 1 where appropriate), one can obtain

$\begin{matrix}{{c_{p} = {\frac{1}{N}\left\lbrack {{a_{1}\left( {1 + ^{- \frac{j2\pi}{r}} + ^{- \frac{j2\pi 2}{r}} + \ldots + ^{- \frac{{j2\pi}{({w - 1})}}{r}}} \right)} + {a_{2}\left( {^{- \frac{{j2\pi}{(r)}}{r}} + ^{- \frac{{j2\pi}{({r + 1})}}{r}} + \ldots + ^{- \frac{{j2\pi}{({r + w - 1})}}{r}}} \right)} + {a_{3}\left( {^{- \frac{{j2\pi}{({2\; r})}}{r}} + ^{- \frac{{j2\pi}{({{2\; r} + 1})}}{r}} + \ldots + ^{- \frac{{j2\pi}{({{2\; r} + w - 1})}}{r}}} \right)} + \ldots + {a_{p}\left( {^{- \frac{{j2\pi}{({{({p - 1})}r})}}{r}} + ^{- \frac{{j2\pi}{({{{({p - 1})}r} + 1})}}{r}} + \ldots + ^{- \frac{{j2\pi}{({{{({p - 1})}r} + w - 1})}}{r}}} \right)}} \right\rbrack}}\mspace{20mu} {c_{k} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1}\; {{x(n)}^{{- {({{j2\pi}\; {kn}})}}/N}}}}}} & (5)\end{matrix}$

One can express everything inside the round brackets as X. Then, c_(p)can be expressed as

$\begin{matrix}{\begin{matrix}{c_{p} = {\frac{1}{N}\left\lbrack {{a_{1}X} + {a_{2}X} + {a_{3}X} + \ldots + {a_{p}X}} \right\rbrack}} \\{= {\frac{X}{N}\left\lbrack {a_{1} + a_{2} + a_{3} + \ldots + a_{p}} \right\rbrack}}\end{matrix}{c_{k} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1}\; {{x(n)}^{{- {({{j2\pi}\; {kn}})}}/N}}}}}} & (6)\end{matrix}$

The foregoing simplification that yields Equation 6 can be performed asfollows. In Equation 4, one can equate the first terms in the curvedbrackets, so that

$\begin{matrix}{{1 = {^{- \frac{{j2\pi}{(r)}}{r}} = {^{- \frac{{j2\pi}{({2\; r})}}{r}} = ^{- \frac{{j2\pi}{({{({p - 1})}r})}}{r}}}}}{c_{k} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1}\; {{x(n)}^{{- {({{j2\pi}\; {kn}})}}/N}}}}}} & (7)\end{matrix}$

Similarly, one can equate the second terms in the curved brackets ofEquation 4 to yield

$\begin{matrix}{{^{- \frac{j2\pi}{r}} = {^{- \frac{{j2\pi}{({r + 1})}}{r}} = {^{- \frac{{j2\pi}{({{2\; r} + 1})}}{r}} = ^{- \frac{{j2\pi}{({{{({p - 1})}r} + 1})}}{r}}}}}{c_{k} = {\frac{1}{N}{\sum\limits_{n = 0}^{N - 1}\; {{x(n)}^{{- {({{j2\pi}\; {kn}})}}/N}}}}}} & (8)\end{matrix}$

Such equating process can be done for all of the terms in the curvedbrackets.

From Equation 6, one can see that as long as the pulse train amplitudesa through a_(p) all sum to zero there is no reference spur. Similarstrategy can be used to determine the spur amplitude for integermultiples of the p^(th) spur (e.g., for any multiple of the referencefrequency).

To demonstrate that Equation 6, is valid or at least sufficient toprovide spur reduction or eradication functionality, MATLAB was used togenerate an arbitrary signal representative of the example of panel (b)in FIG. 14. Such a generated signal is shown in FIG. 15, where the pulsefrequency is at 10 MHz and the pulse widths are 10%.

FIG. 16 shows the frequency domain representation of the time domainsignal of FIG. 15. It is noted that the nulls (294) occur every 10 MHzas predicted by Equation 6. It is also noted that the sync functionshape has a null (292) at 100 MHz. Having an example 10% pulse widthyields the first sync null at 10 times the reference. Having an examplepulse width of 100% would put the sync pulse null on top of the firstreference pulse null. This implies that in some situations, the firstreference frequency is the furthest the sync pulse null can be pulledin.

The foregoing example demonstrates two methods in which reference spurscan be eliminated. First, nulls can occur at integer multiples of thereference frequency. Second, nulls can be arbitrarily placed at anyfrequency greater than the reference frequency by changing the pulsewidth.

FIG. 17 shows an example configuration 300 that can be implemented toprovide one or more functionalities as described herein. The exampleconfiguration 300 is described as a fixed width variable amplitudecharge pump (FWVACP). FIG. 18 shows timing diagrams for variousparameters associated with the FWVACP configuration of FIG. 17.

A phase detector 172 is shown to have as inputs reference signal(frequency f_(ref)) and feedback signal (frequency f_(fb)) in a PLL. Forthe purpose of description, these signals can also be referred to bytheir respective frequencies—for example, f_(ref) for the referenceclock signal, and f_(fb) for the feedback signal.

In the phase detector 172, f_(ref) is compared with f_(fb). Based on thecomparison, the phase detector 172 can generate an ‘up’ or ‘down’ pulsethat relates to the phase error for a given cycle of f_(ref). Suchcomparison of the input signals and generation of the ‘up’ or ‘down’pulse can be achieved by, for example, a tristate phase detectorconfiguration.

The ‘up’ and ‘down’ pulses can be OR'ed (302) together to generate a‘charge_switch’ signal which is provided to and engage a charging switch308. With the charging switch 308 engaged, current I₁ (from a currentsource 304 and through path 306) can charge a capacitor C. The charge onthe capacitor C does not know if it resulted from an ‘up’ or ‘down’pulse. Such information can be recorded in the signals from the ‘UpPulse’ and ‘Down Pulse’ blocks, described below in greater detail.

The charge on the capacitor C represents the accumulated phase error fora single reference clock cycle as defined by clock signal f_(ref). A‘Voltage to Current’ circuit can be configured to receive (through path310) and convert the voltage over C into a current I₂ generated by acurrent source 322 or 328 (through their respective control paths 318and 320). With the current I₂ now set, a set of switches can be providedand configured to pass the current on to a loop filter 180. For example,an ‘Up Pulse Switch’ 324 under the control of an ‘Up Pulse’ block 330(with switching signal ‘up_pulse_on’) can allow positive I₂ to beprovided from the current source 322 to the loop filter 180. Similarly,a ‘Down Pulse Switch’ 326 under the control of a ‘Down Pulse’ block 332(with switching signal ‘dn_pulse_on’) can allow negative I₂ to beprovided from the current source 328 to the loop filter 180.

It is noted that an output signal that is generated by the FWVACPpreferably has a fixed width. In some implementations, such an outputcan be facilitated by configuring the ‘Up Pulse’ block 330 and the ‘DownPulse’ block 332 so that their respective output signals ‘up_pulse_on’and ‘dn_pulse_on’ include pulses that have a substantially constantwidth.

In some implementations, such a constant width of the ‘up_pulse_on’ and‘dn_pulse_on’ signals can be controlled by a ‘pulse’ signal shown to beinput into each of the ‘Up Pulse’ block 330 and the ‘Down Pulse’ block332. For example, and as shown in the timing diagrams of FIG. 18, the‘pulse’ signal can begin on the falling edge (352) of f_(ref) in orderto capture phase error information from both ‘up’ and ‘down’ pulses.

Based on the foregoing ‘pulse’ signal as an input, the ‘Up Pulse’ block330 can be configured to operate as follows. Signals ‘up’ and ‘pulse’are fed into the block (330). The ‘up’ signal can be obtained from theoutput of the phase detector 172 as described herein. A signal generatedwithin the ‘Up Pulse’ block 330 and referred to herein as ‘up_toggle’can be configured to go high when an ‘up’ rising edge (350 in FIG. 18)is emitted from the phase detector 172, and go low at the falling edge(e.g., the first falling edge) of the ‘pulse’ signal. The ‘up_toggle’signal and the ‘pulse’ signal can be combined (e.g., an AND operation)to generate the ‘up_pulse_on’ signal. As described herein, the‘up_pulse_on’ can be provided to and engage the ‘Up Pulse Switch’ 324.

Similarly, the ‘Down Pulse’ block 332 can be configured to operate basedon the ‘pulse’ signal as an input. Signals ‘dn’ and ‘pulse’ are fed intothe block (332). The ‘dn’ signal can be obtained from the output of thephase detector 172 as described herein. A signal generated within the‘Down Pulse’ block 332 and referred to herein as ‘dn_toggle’ can beconfigured to go high when a ‘dn’ rising edge is emitted from the phasedetector 172, and go low at the falling edge (e.g., the first fallingedge) of the ‘pulse’ signal. The ‘dn_toggle’ signal and the ‘pulse’signal can be combined (e.g., an AND operation) to generate the‘dn_pulse_on’ signal. As described herein, the ‘dn_pulse_on’ can beprovided to and engage the ‘Down Pulse Switch’ 326.

As shown in FIG. 17, with either the ‘Up Pulse Switch’ 324 or the ‘DownPulse Switch’ 326 engaged, a respective current is passed on to the loopfilter 180. As further shown in FIGS. 17 and 18, a ‘dump_cap’ signal canbe provided (through path 314) to a ‘Drain Cap Switch’ 312 to allowdraining of the charge on C at an appropriate time by engaging the‘Drain Cap Switch’ 312.

In the foregoing example of the FWVACP operation, timing of the varioussignals can be important. For example, it is preferable to have the ‘UpPulse Switch’ 324 or the ‘Down Pulse Switch’ 326 be engaged when thecapacitor C has substantially fully stored the charge for a givenreference cycle. In some implementations, the ‘Up Pulse Switch’ 324 orthe ‘Down Pulse Switch’ 326 is only engaged when the capacitor C hasfully stored the charge for a given reference cycle.

Said another way, it is preferable that if either of the ‘Up PulseSwitch’ 324 or the ‘Down Pulse Switch’ 326 is engaged, the ‘ChargingSwitch’ 308 not be engaged. In some implementations, if either of the‘Up Pulse Switch’ 324 or the ‘Down Pulse Switch’ 326 is engaged, the‘Charging Switch’ 308 is not engaged.

It is preferable to have the capacitor C be substantially fully drainedbefore the ‘Charging Switch’ 308 is engaged. In some implementations,the capacitor C is fully drained before the ‘Charging Switch’ 308 isengaged.

It is preferable that the capacitor C not be draining if an output pulseis being provided to the loop filter 180. In some implementations, thecapacitor C is not draining if an output pulse is being provided to theloop filter 180.

As described herein, one or more features of the present disclosurerelates to a technique for eliminating reference spurs in a Frac-N PLL.Such a technique can be implemented by manipulating the charge pumppulse in such a way that it has a substantially constant width andoccurs periodically. Phase error information can be embedded in theamplitude of the pulse. In some implementations, additional frequenciesoutside the first reference frequency may also be targeted by settingthe width of the pulse.

The present disclosure describes various features, no single one ofwhich is solely responsible for the benefits described herein. It willbe understood that various features described herein may be combined,modified, or omitted, as would be apparent to one of ordinary skill.Other combinations and sub-combinations than those specificallydescribed herein will be apparent to one of ordinary skill, and areintended to form a part of this disclosure. Various methods aredescribed herein in connection with various flowchart steps and/orphases. It will be understood that in many cases, certain steps and/orphases may be combined together such that multiple steps and/or phasesshown in the flowcharts can be performed as a single step and/or phase.Also, certain steps and/or phases can be broken into additionalsub-components to be performed separately. In some instances, the orderof the steps and/or phases can be rearranged and certain steps and/orphases may be omitted entirely. Also, the methods described herein areto be understood to be open-ended, such that additional steps and/orphases to those shown and described herein can also be performed.

Some aspects of the systems and methods described herein canadvantageously be implemented using, for example, computer software,hardware, firmware, or any combination of computer software, hardware,and firmware. Computer software can comprise computer executable codestored in a computer readable medium (e.g., non-transitory computerreadable medium) that, when executed, performs the functions describedherein. In some embodiments, computer-executable code is executed by oneor more general purpose computer processors. A skilled artisan willappreciate, in light of this disclosure, that any feature or functionthat can be implemented using software to be executed on a generalpurpose computer can also be implemented using a different combinationof hardware, software, or firmware. For example, such a module can beimplemented completely in hardware using a combination of integratedcircuits. Alternatively or additionally, such a feature or function canbe implemented completely or partially using specialized computersdesigned to perform the particular functions described herein ratherthan by general purpose computers.

Multiple distributed computing devices can be substituted for any onecomputing device described herein. In such distributed embodiments, thefunctions of the one computing device are distributed (e.g., over anetwork) such that some functions are performed on each of thedistributed computing devices.

Some embodiments may be described with reference to equations,algorithms, and/or flowchart illustrations. These methods may beimplemented using computer program instructions executable on one ormore computers. These methods may also be implemented as computerprogram products either separately, or as a component of an apparatus orsystem. In this regard, each equation, algorithm, block, or step of aflowchart, and combinations thereof, may be implemented by hardware,firmware, and/or software including one or more computer programinstructions embodied in computer-readable program code logic. As willbe appreciated, any such computer program instructions may be loadedonto one or more computers, including without limitation a generalpurpose computer or special purpose computer, or other programmableprocessing apparatus to produce a machine, such that the computerprogram instructions which execute on the computer(s) or otherprogrammable processing device(s) implement the functions specified inthe equations, algorithms, and/or flowcharts. It will also be understoodthat each equation, algorithm, and/or block in flowchart illustrations,and combinations thereof, may be implemented by special purposehardware-based computer systems which perform the specified functions orsteps, or combinations of special purpose hardware and computer-readableprogram code logic means.

Furthermore, computer program instructions, such as embodied incomputer-readable program code logic, may also be stored in a computerreadable memory (e.g., a non-transitory computer readable medium) thatcan direct one or more computers or other programmable processingdevices to function in a particular manner, such that the instructionsstored in the computer-readable memory implement the function(s)specified in the block(s) of the flowchart(s). The computer programinstructions may also be loaded onto one or more computers or otherprogrammable computing devices to cause a series of operational steps tobe performed on the one or more computers or other programmablecomputing devices to produce a computer-implemented process such thatthe instructions which execute on the computer or other programmableprocessing apparatus provide steps for implementing the functionsspecified in the equation(s), algorithm(s), and/or block(s) of theflowchart(s).

Some or all of the methods and tasks described herein may be performedand fully automated by a computer system. The computer system may, insome cases, include multiple distinct computers or computing devices(e.g., physical servers, workstations, storage arrays, etc.) thatcommunicate and interoperate over a network to perform the describedfunctions. Each such computing device typically includes a processor (ormultiple processors) that executes program instructions or modulesstored in a memory or other non-transitory computer-readable storagemedium or device. The various functions disclosed herein may be embodiedin such program instructions, although some or all of the disclosedfunctions may alternatively be implemented in application-specificcircuitry (e.g., ASICs or FPGAs) of the computer system. Where thecomputer system includes multiple computing devices, these devices may,but need not, be co-located. The results of the disclosed methods andtasks may be persistently stored by transforming physical storagedevices, such as solid state memory chips and/or magnetic disks, into adifferent state.

Unless the context clearly requires otherwise, throughout thedescription and the claims, the words “comprise,” “comprising,” and thelike are to be construed in an inclusive sense, as opposed to anexclusive or exhaustive sense; that is to say, in the sense of“including, but not limited to.” The word “coupled”, as generally usedherein, refers to two or more elements that may be either directlyconnected, or connected by way of one or more intermediate elements.Additionally, the words “herein,” “above,” “below,” and words of similarimport, when used in this application, shall refer to this applicationas a whole and not to any particular portions of this application. Wherethe context permits, words in the above Detailed Description using thesingular or plural number may also include the plural or singular numberrespectively. The word “or” in reference to a list of two or more items,that word covers all of the following interpretations of the word: anyof the items in the list, all of the items in the list, and anycombination of the items in the list. The word “exemplary” is usedexclusively herein to mean “serving as an example, instance, orillustration.” Any implementation described herein as “exemplary” is notnecessarily to be construed as preferred or advantageous over otherimplementations.

The disclosure is not intended to be limited to the implementationsshown herein. Various modifications to the implementations described inthis disclosure may be readily apparent to those skilled in the art, andthe generic principles defined herein may be applied to otherimplementations without departing from the spirit or scope of thisdisclosure. The teachings of the invention provided herein can beapplied to other methods and systems, and are not limited to the methodsand systems described above, and elements and acts of the variousembodiments described above can be combined to provide furtherembodiments. Accordingly, the novel methods and systems described hereinmay be embodied in a variety of other forms; furthermore, variousomissions, substitutions and changes in the form of the methods andsystems described herein may be made without departing from the spiritof the disclosure. The accompanying claims and their equivalents areintended to cover such forms or modifications as would fall within thescope and spirit of the disclosure.

1. (canceled)
 2. A circuit for a frequency synthesizer of a wirelessdevice, comprising: a phase frequency detector (PFD) configured togenerate a first signal based on a reference signal and a feedbacksignal; and a compensation circuit configured to generate a compensationsignal based on the first signal, the compensation circuit including acharging circuit, the charging circuit including a capacitance element,a charging switch configured to facilitate charging of the capacitanceelement in response to the first signal, and a drain switch configuredto facilitate draining of the charge in the capacitance element.
 3. Thecircuit of claim 2 wherein the compensation circuit further includes acharge pump configured to generate a current signal to compensate for aphase difference between the reference signal and the feedback signal.4. The circuit of claim 3 wherein the compensation circuit furtherincludes a voltage-to-current converter configured to generate a controlsignal in response to a voltage of the capacitance element, the chargepump configured to generate the current signal based on the controlsignal from the voltage-to-current converter.
 5. The circuit of claim 3wherein the compensation circuit further includes a loop filterconfigured to receive the current signal and generate a correspondingvoltage signal, the voltage signal being provided to a voltagecontrolled oscillator (VCO) in communication with the compensationcircuit.
 6. The circuit of claim 3 wherein the compensation signalincludes the current signal having a pulse with a substantially constantwidth and an amplitude representative of the phase difference.
 7. Thecircuit of claim 3 wherein the PFD is configured to generate an upsignal or a down (dn) signal as the first signal, the up signal beinggenerated when a phase of the reference signal leads a phase of thefeedback signal, the dn signal being generated when a phase of thereference signal lags a phase of the feedback signal.
 8. The circuit ofclaim 7 wherein the charge pump includes a first current sourceconfigured to generate a positive current as the current signal for theup signal and a second current source configured to generate a negativecurrent as the current signal for the dn signal.
 9. The circuit of claim7 wherein the current signal is modulated based on a combination of theup signal and a pulse signal or a combination of the dn signal and thepulse signal.
 10. The circuit of claim 7 wherein the compensationcircuit further includes a first switch in communication with a firstcurrent source configured to generate a positive current as the currentsignal for the up signal and a second switch in communication with asecond current source configured to generate a negative current as thecurrent signal for the dn signal, the first and second switchesconfigured to be controlled to provide a modulation of the currentsignal.
 11. The circuit of claim 7 wherein the compensation circuitfurther includes a first switch configured to cause a positive currentto be generated as the current signal and a second switch configured tocause a negative current to be generated as the current signal, thefirst switch or the second switch being engaged only when thecapacitance element has a substantially full charge.
 12. The circuit ofclaim 7 wherein the compensation circuit further includes a first switchconfigured to cause a positive current to be generated as the currentsignal and a second switch configured to cause a negative current to begenerated as the current signal, the charging switch not being engagedif either of the first switch and the second switch is engaged.
 13. Thecircuit of claim 7 wherein the compensation circuit further includes afirst control block in communication with a first switch and a secondcontrol block in communication with a second switch, the first controlblock configured to generate a first enable pulse for the first switchbased on a combination of the up signal and a pulse signal, the secondcontrol block configured to generate a second enable pulse for thesecond switch based on a combination of the dn signal and the pulsesignal.
 14. The circuit of claim 7 wherein the compensation circuitfurther includes a first control block configured to generate a firstenable signal based on the pulse signal and a first internal signal thatgoes high with a rising edge of the up signal and low with a fallingedge of the pulse signal and a second control block configured togenerate a second enable signal based on the pulse signal and a secondinternal signal that goes high with a rising edge of the dn signal andlow with the falling edge of the pulse signal.
 15. The circuit of claim7 wherein the current signal is modulated based on a combination of theup signal and a pulse signal or a combination of the dn signal and thepulse signal, the pulse signal including a plurality of pulses, eachpulse having a substantially constant width and going high with afalling edge of the reference signal.
 16. The circuit of claim 2 furthercomprising a voltage-controlled oscillator (VCO) in communication withthe compensation circuit, the VCO configured to generate an outputsignal based on the compensation signal.
 17. The circuit of claim 16further comprising a divider circuit in communication with the VCO andthe PFD, the divider circuit configured to receive the output signalfrom the VCO and generate an updated version of the feedback signal. 18.The circuit of claim 17 further comprising a sigma delta modulator (SDM)in communication with the divider circuit to form a loop, the loop beingconfigured to allow the output signal to have an output frequency thatis a non-integer multiple of the frequency of the reference signal. 19.The circuit of claim 2 wherein the circuit is a Frac-N phase-locked loop(PLL) circuit.
 20. A method for operating a frequency synthesizer of awireless device, comprising: generating a first signal based on areference signal and a feedback signal; and with a compensation circuitincluding a charging circuit including a capacitance element, a chargingswitch configured to facilitate charging of the capacitance element inresponse to the first signal, and a drain switch configured tofacilitate draining of the charge in the capacitance element, generatinga compensation signal based on the generated first signal.
 21. Awireless communication device, comprising: an antenna configured tofacilitate reception of a radio-frequency (RF) signal; a receiver incommunication with the antenna, the receiver configured to process theRF signal; and a frequency synthesizer circuit in communication with thereceiver, the frequency synthesizer circuit including a phase frequencydetector (PFD) configured to generate a first signal based on areference signal and a feedback signal, the frequency synthesizercircuit further including a compensation circuit configured to generatea compensation signal based on the first signal, the compensationcircuit including a charging circuit including a capacitance element, acharging switch configured to facilitate charging of the capacitanceelement in response to the first signal, and a drain switch configuredto facilitate draining of the charge in the capacitance element.